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Philips Semiconductors RF Communications Products
Product specification
Programmable analog compandor
NE/SA572
DESCRIPTION
The NE572 is a dual-channel, high-performance gain control circuit in which either channel may be used for dynamic range compression or expansion. Each channel has a full-wave rectifier to detect the average value of input signal, a linearized, temperature-compensated variable gain cell (G) and a dynamic time constant buffer. The buffer permits independent control of dynamic attack and recovery time with minimum external components and improved low frequency gain control ripple distortion over previous compandors. The NE572 is intended for noise reduction in high-performance audio systems. It can also be used in a wide range of communication systems and video recording applications.
FEATURES
time
* Independent control of attack and recovery * Improved low frequency gain control ripple * Complementary gain compression and
expansion with external op amp
PIN CONFIGURATION
D1, N, F Packages
TRACK TRIM A 1 RECOV. CAP A 2 RECT. IN A 3 ATTACK CAP A 4 G OUT A 5 THD TRIM A 6 G IN A 7 GND 8 16 VCC
15 TRACK TRIM B 14 RECOV. CAP B 13 RECT. IN B 12 ATTACK CAP B 11 G OUT B 10 THD TRIM B 9 G IN B
* Wide dynamic range--greater than 110dB * Temperature-compensated gain control * Low distortion gain cell * Low noise--6V typical * Wide supply voltage range--6V-22V * System level adjustable with external
components
NOTE: 1. D package released in large SO (SOL) package only.
APPLICATIONS
* Dynamic noise reduction system * Voltage control amplifier * Stereo expandor * Automatic level control * High-level limiter * Low-level noise gate * State variable filter
ORDERING INFORMATION
DESCRIPTION 16-Pin Plastic Small Outline (SO) 16-Pin Plastic Dual In-Line Package (DIP) 16-Pin Plastic Small Outline (SO) 16-Pin Ceramic Dual In-Line Package (Cerdip) 16-Pin Plastic Dual In-Line Package (DIP) TEMPERATURE RANGE 0 to +70C 0 to +70C -40 to +85C -40 to +85C -40 to +85C ORDER CODE NE572D NE572N SA572D SA572F SA572N DWG # 0005 0406 0005 0582 0406
ABSOLUTE MAXIMUM RATINGS
SYMBOL VCC TA PARAMETER Supply voltage Operating temperature range NE572 SA572 PD Power dissipation 0 to +70 -40 to +85 500 mW C RATING 22 UNIT VDC
October 7, 1987
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853-0813 90829
Philips Semiconductors RF Communications Products
Product specification
Programmable analog compandor
NE/SA572
BLOCK DIAGRAM
R1 (7,9) 6.8k (6,10) 500 GAIN CELL G (5,11)
(1,15) (3,13) - + 270 10k RECTIFIER - + BUFFER 10k
(16)
P.S.
(8)
(4,12)
(2,14)
DC ELECTRICAL CHARACTERISTICS
Standard test conditions (unless otherwise noted) VCC=15V, TA=25C; Expandor mode (see Test Circuit). Input signals at unity gain level (0dB) = 100mVRMS at 1kHz; V1 = V2; R2 = 3.3k; R3 = 17.3k. SYMBOL VCC ICC VR THD THD THD PARAMETER Supply voltage Supply current Internal voltage reference Total harmonic distortion (untrimmed) Total harmonic distortion (trimmed) Total harmonic distortion (trimmed) No signal output noise DC level shift (untrimmed) Unity gain level Large-signal distortion Tracking error (measured relative to value at unity gain)= [VO-VO (unity gain)]dB -V2dB Channel crosstalk PSRR Power supply rejection ratio 200mVRMS into channel A, measured output on channel B 120Hz 60 70 Rectifier input V2=+6dB V1=0dB V2=-30dB V1=0dB 0.2 0.5 -1.5 +0.8 60 70 0.2 0.5 -2.5 +1.6 dB dB dB V1=V2=400mV 1kHz CA=1.0F 1kHz CR=10F 100Hz Input to V1 and V2 grounded (20-20kHz) Input change from no signal to 100mVRMS -1 No signal 2.3 2.5 0.2 0.05 0.25 6 20 0 0.7 25 50 +1 3.0 -1.5 TEST CONDITIONS Min 6 NE572 Typ Max 22 6 2.7 1.0 2.3 2.5 0.2 0.05 0.25 6 20 0 0.7 25 50 +1.5 3 Min 6 SA572 Typ Max 22 6.3 2.7 1.0 VDC mA VDC % % % V mV dB % UNIT
October 7, 1987
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Philips Semiconductors RF Communications Products
Product specification
Programmable analog compandor
NE/SA572
TEST CIRCUIT
1F 22F 2.2F V1 (7,9) 6.8k G (5,11) 82k 1% R3 17.3k + 100 -15V
- 5 = 10F BUFFER 1k (4,12) (8) + 2.2F (2,14) 2.2k (6,10) + 270pF NE5234 V0
(1,15)
2.2F V2 R2 1%
3.3k
(3,13) RECTIFIER (16) + .1F 22F
+15V
AUDIO SIGNAL PROCESSING IC COMBINES VCA AND FAST ATTACK/SLOW RECOVERY LEVEL SENSOR
In high-performance audio gain control applications, it is desirable to independently control the attack and recovery time of the gain control signal. This is true, for example, in compandor applications for noise reduction. In high end systems the input signal is usually split into two or more frequency bands to optimize the dynamic behavior for each band. This reduces low frequency distortion due to control signal ripple, phase distortion, high frequency channel overload and noise modulation. Because of the expense in hardware, multiple band signal processing up to now was limited to professional audio applications. With the introduction of the Signetics NE572 this high-performance noise reduction concept becomes feasible for consumer hi fi applications. The NE572 is a dual channel gain control IC. Each channel has a linearized, temperature-compensated gain cell and an improved level sensor. In conjunction with an external low noise op amp for current-to-voltage conversion, the VCA features low distortion, low noise and wide dynamic range. The novel level sensor which provides gain control current for the VCA gives lower gain control ripple and independent control of fast attack, slow recovery dynamic response. An
attack capacitor CA with an internal 10k resistor RA defines the attack time tA. The recovery time tR of a tone burst is defined by a recovery capacitor CR and an internal 10k resistor RR. Typical attack time of 4ms for the high-frequency spectrum and 40ms for the low frequency band can be obtained with 0.1F and 1.0F attack capacitors, respectively. Recovery time of 200ms can be obtained with a 4.7F recovery capacitor for a 100Hz signal, the third harmonic distortion is improved by more than 10dB over the simple RC ripple filter with a single 1.0F attack and recovery capacitor, while the attack time remains the same. The NE572 is assembled in a standard 16-pin dual in-line plastic package and in oversized SOL package. It operates over a wide supply range from 6V to 22V. Supply current is less than 6mA. The NE572 is designed for consumer application over a temperature range 0-70 The SA572 is intended for applications from -40C to +85C.
inherent low distortion, low noise and the capability to linearize large signals, a wide dynamic range can be obtained. The buffer amplifiers are provided to permit control of attack time and recovery time independent of each other. Partitioned as shown in the block diagram, the IC allows flexibility in the design of system levels that optimize DC shift, ripple distortion, tracking accuracy and noise floor for a wide range of application requirements.
Gain Cell
Figure 1 shows the circuit configuration of the gain cell. Bases of the differential pairs Q1-Q2 and Q3-Q4 are both tied to the output and inputs of OPA A1. The negative feedback through Q1 holds the VBE of Q1-Q2 and the VBE of Q3-Q4 equal. The following relationship can be derived from the transistor model equation in the forward active region. DV BE
Q3Q4
+ D BE
Q1Q2
NE572 BASIC APPLICATIONS Description
The NE572 consists of two linearized, temperature-compensated gain cells (G), each with a full-wave rectifier and a buffer amplifier as shown in the block diagram. The two channels share a 2.5V common bias reference derived from the power supply but otherwise operate independently. Because of
(VBE = VT IIN IC/IS)
October 7, 1987
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Philips Semiconductors RF Communications Products
Product specification
Programmable analog compandor
NE/SA572
V TI n
1 I) 2G
1 I 2O
IS V IN R1
*
V TI n
1 I* 2G
1 I 2O
V+
IS
where I IN +
R1 = 6.8k I1 = 140A I2 = 280A
V TI n
I1 ) IS
I IN
*
V TI n
I2 *
I1 * IS
I IN
1 I) 2G
1 I 2O
(2)
A1 + Q3 -
I1 140A
where I IN +
V IN R1
IO
R1 = 6.8k I1 = 140A I2 = 280A IO is the differential output current of the gain cell and IG is the gain control current of the gain cell. If all transistors Q1 through Q4 are of the same size, equation (2) can be simplified to: 2 + @ I2 1 I* I2 2
Q4
Q1
Q2
R1 6.8k IG THD TRIM VREF I2 280A
VIN
IO
I IN @
IG *
2I 1 @
IG
(3)
Figure 1. Basic Gain Cell Schematic The internal bias scheme limits the maximum output current IR to be around 300A. Within a 1dB error band the input range of the rectifier is about 52dB.
The first term of Equation 3 shows the multiplier relationship of a linearized two quadrant transconductance amplifier. The second term is the gain control feedthrough due to the mismatch of devices. In the design, this has been minimized by large matched devices and careful layout. Offset voltage is caused by the device mismatch and it leads to even harmonic distortion. The offset voltage can be trimmed out by feeding a current source within 25A into the THD trim pin. The residual distortion is third harmonic distortion and is caused by gain control ripple. In a compandor system, available control of fast attack and slow recovery improve ripple distortion significantly. At the unity gain level of 100mV, the gain cell gives THD (total harmonic distortion) of 0.17% typ. Output noise with no input signals is only 6V in the audio spectrum (10Hz-20kHz). The output current IO must feed the virtual ground input of an operational amplifier with a resistor from output to inverting input. The non-inverting input of the operational amplifier has to be biased at VREF if the output current IO is DC coupled.
Rectifier
The rectifier is a full-wave design as shown in Figure 2. The input voltage is converted to current through the input resistor R2 and turns on either Q5 or Q6 depending on the signal polarity. Deadband of the voltage to current converter is reduced by the loop gain of the gain block A2. If AC coupling is used, the rectifier error comes only from input bias current of gain block A2. The input bias current is typically about 70nA. Frequency response of the gain block A2 also causes second-order error at high frequency. The collector current of Q6 is mirrored and summed at the collector of Q5 to form the full wave rectified output current IR. The rectifier transfer function is
IR +
V IN * R2
V REF
(4)
If VIN is AC-coupled, then the equation will be reduced to:
I RAC +
V IN(AVG) R2
October 7, 1987
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Philips Semiconductors RF Communications Products
Product specification
Programmable analog compandor
NE/SA572
V+
IR +
V IN *
V REF R2
Buffer Amplifier
In audio systems, it is desirable to have fast attack time and slow recovery time for a tone burst input. The fast attack time reduces transient channel overload but also causes low-frequency ripple distortion. The low-frequency ripple distortion can be improved with the slow recovery time. If different attack times are implemented in corresponding frequency spectrums in a split band audio system, high quality performance can be achieved. The buffer amplifier is designed to make this feature available with minimum external components. Referring to Figure 3, the rectifier output current is mirrored into the input and output of the unipolar buffer amplifier A3 through Q8, Q9 and Q10. Diodes D11 and D12 improve tracking accuracy and provide common-mode bias for A3. For a positive-going input signal, the buffer amplifier acts like a voltage-follower. Therefore, the output impedance of A3 makes the contribution of capacitor CR to attack time insignificant. Neglecting diode impedance, the gain Ga(t) for G can be expressed as follows: Ga(t) + (Ga INT * Ga FNL e t
* t A
VREF
+ A2 - Q5
D7
Q6 R2 VIN
Figure 2. Simplified Rectifier Schematic
)
Ga FNL
V+ Q8 Q9 Q10
GaINT=Initial Gain GaFNL=Final Gain A=RA * CA=10k * CA
IQ
= 2IR2
Q17
IR2 X2 Q16
IR +
V IN R
- A3 + 10k IR1 D15
10k
where A is the attack time constant and RA is a 10k internal resistor. Diode D15 opens the feedback loop of A3 for a negative-going signal if the value of capacitor CR is larger than capacitor CA. The recovery time depends only on CR * RR. If the diode impedance is assumed negligible, the dynamic gain GR (t) for G is expressed as follows.
D13
G R(t) + (G RINT *
G RFNL e t
*
t R
)
G RFNL
GR(t)=(GR INT-GR FNL) e +GR FNL
Q14 X2 Q18
R=RR * CR=10k * CR where R is the recovery time constant and RR is a 10k internal resistor. The gain control current is mirrored to the gain cell through Q14. The low level gain errors due to input bias current of A2 and A3 can be trimmed through the tracking trim pin into A3 with a current source of 3A.
D11 D12
CR CA TRACKING TRIM
Basic Expandor
Figure 4 shows an application of the circuit as a simple expandor. The gain expression of the system is given by
Figure 3. Buffer Amplifier Schematic
October 7, 1987
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Philips Semiconductors RF Communications Products
Product specification
Programmable analog compandor
NE/SA572
V OUT 2 + @ V IN I1
(I1=140A)
R 3 @ V IN(AVG) R2 @ R1
(5)
Both the resistors R1 and R2 are tied to internal summing nodes. R1 is a 6.8k internal resistor. The maximum input current into the gain cell can be as large as 140A. This corresponds to a voltage level of 140A * 6.8k=952mV peak. The input peak current into the rectifier is limited to 300A by the internal bias system. Note that the value of R1 can be increased to accommodate higher input level. R2 and R3 are external resistors. It is easy to adjust the ratio of R3/R2 for
desirable system voltage and current levels. A small R2 results in higher gain control current and smaller static and dynamic tracking error. However, an impedance buffer A1 may be necessary if the input is voltage drive with large source impedance. The gain cell output current feeds the summing node of the external OPA A2. R3 and A2 convert the gain cell output current to the output voltage. In high-performance applications, A2 has to be low-noise, high-speed and wide band so that the high-performance output of the gain cell will not be degraded. The non-inverting input of A2 can be biased at the low noise internal
reference Pin 6 or 10. Resistor R4 is used to bias up the output DC level of A2 for maximum swing. The output DC level of A2 is given by
V ODC + V REF 1 )
R3 R4
*
VB
R3 R4
(6)
VB can be tied to a regulated power supply for a dual supply system and be grounded for a single supply system. CA sets the attack time constant and CR sets the recovery time constant. *5COL
R4 +VB
R3 17.3k
- CIN1 VIN 2.2F + A1
CIN2 (7,9)
R1 6.8k
G
(5,11) A2 (6,10) R6 VREF 1k (2,14) VOUT
R5 100k
BUFFER CIN3 2.2F R2 3.3k (3,13)
(4,12)
C1 2.2F
CA CR 1F 10F
(8)
(16) +VCC
Figure 4. Basic Expandor Schematic
October 7, 1987
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Philips Semiconductors RF Communications Products
Product specification
Programmable analog compandor
NE/SA572
Basic Compressor
Figure 5 shows the hook-up of the circuit as a compressor. The IC is put in the feedback loop of the OPA A1. The system gain expression is as follows:
R4
RDC1 9.1k CDC 10F C2 .1F
RDC2 9.1k
V OUT + V IN
I1 R2 @ R1 @ 2 R 3 @ V IN(AVG)
1 2
(7)
VIN
CIN1 - 2.2F R3 17.3k + C1 1k R5 (6,10) VREF R1 6.8k A1
D1
D2
RDC1, RDC2, and CDC form a DC feedback for A1. The output DC level of A1 is given by
VOUT
V ODC + V REF 1 )
*
R DC1 ) R DC2 R4
(8)
VB @
R DC1 ) R DC2 R4
(5,11) (2,14)
G
(7,9) CIN2 2.2F
The zener diodes D1 and D2 are used for channel overload protection.
Basic Compandor System
The above basic compressor and expandor can be applied to systems such as tape/disc noise reduction, digital audio, bucket brigade delay lines. Additional system design techniques such as bandlimiting, band splitting, pre-emphasis, de-emphasis and equalization are easy to incorporate. The IC is a versatile functional block to achieve a high performance audio system. Figure 6 shows the system level diagram for reference.
(4,12)
BUFFER
CIN3 2.2F
3.3k R2 CR 10F CA 1F (8) VCC (16) (3,13)
Figure 5. Basic Compressor Schematic
October 7, 1987
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Philips Semiconductors RF Communications Products
Product specification
Programmable analog compandor
NE/SA572
1 2 VRMS COMPRESSION IN INPUT TO G AND RECT
2 REL LEVEL EXPANDOR OUT +29.54 dB ABS LEVEL dBM
3.0V
+11.76
547.6MV 400MV
+14.77 +12.0
-3.00 -5.78
100MV
0.0
-17.78
10MV
-20
-37.78
1MV
-40
-57.78
100V
-60
-77.78
10V
-80
-97.78
Figure 6. NE572 System Level
October 7, 1987
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